1. Field of the Invention
The present invention relates to multi-carrier modulation receivers, in particular to a method of canceling the inter-symbol and inter-carrier interface in a multi-carrier modulation receiver.
2. Description of Related Art
MCM (multi-carrier modulation) is a modulation scheme that is widely used in high-speed data communications. It has two flavors: OFDM (orthogonal frequency division multiplexing), which is currently used in wireless local area network standards IEEE 802.11a and 802.11g, and DMT (discrete multi-tone), which is currently used in ADSL (asymmetrical digital subscriber line) standards. While the DMT system is discussed throughout the specification and is used as an example, one of ordinary skill in the art will realize that the techniques disclosed by the present invention are applicable to a general MCM system.
A DMT transmitter utilizes a plurality of tones, (or so-called sub-carriers, which are sinusoidal waves), which are orthogonal to one another. Each tone may carry a certain bit-load of information using a certain modulation scheme, such as 4-QAM (4-point quadrature amplitude modulation, which carries 2-bit information), 16-QAM (16-point quadrature amplitude modulation, which carries 4-bit information), 64-QAM (64-point quadrature amplitude modulation, which carries 6-bit information), and so on. The total bit loads provided by all the tones determines the total number of data bits that a DMT symbol carries. For example, if there are 255 tones and each of them uses 16-QAM, then the total amount of data a DMT symbol carries is 255×4=1020 bits. A DMT transmission system thus operates on a per-frame basis. Each frame consists of a block of data bit stream whose length is equal to the total number of bits that a DMT symbol carries. For example, if each DMT symbol carries 1020 bits, then the data to be transmitted from the transmitter are divided into many blocks; each block has 1020 bits and is carried by a DMT symbol.
FIG. 1 depicts a typical block diagram of the physical layer implementation of a DMT transmitter 1000. A SIPO (serial-in parallel-out) buffer 1100 converts a block of data bit stream (that a DMT symbol carries) into N parallel data banks, labeled 0, 1, 2, to N−1. Each entry of the N banks is mapped into a respective QAM (quadrature amplitude modulation) constellation point by the subsequent QAM mapper 1200, resulting in Ncomplex (frequency domain) data, labeled 0, 1, 2, to N−1. A constellation diagram is a representation of a digital modulation scheme in the complex plane. The points on a constellation diagram are called constellation points. Constellation points are a set of modulation symbols that comprise a modulation alphabet.
For example, if tone number 5 uses 4-QAM to carry two bits of information, the QAM mapper 1200 will map the two-bit input corresponding to tone number 5 into one of the four constellation points: 1+j, 1−j, −1+j, −1−j. The N complex data from QAM mapper 1200 are converted into N time-domain samples by N-point IFFT (inverse Fast Fourier Transform) 1300. The last N_CP time-domain samples of the IFFT output are pre-pended to the beginning of the N samples, resulting in N+N_CP time-domain samples, labeled 0, 1, 2, to N+N_CP−1. These N_CP pre-pended samples are referred to as “cyclic prefix” (CP) of the corresponding IFFT frame. A subsequent PISO (parallel-in serial-out) buffer 1500 converts the N+N_CP time domain samples into N+N_CP serial samples, which is converted into analog voltage by a DAC (digital-analog converter) 1600. An amplifier 1700 amplifies the output from DAC 1600 to make it suitable for transmission over the communication medium 2020 (for example, telephone lines). The N+N_CP serial samples constitute a DMT symbol that carries a certain block of information.
FIG. 2 illustrates the construction of two successive DMT symbols. The last N_CP samples of the IFFT output corresponding to the first block of data bit stream are pre-pended to the beginning of the first DMT symbol. Similarly, the last N_CP samples of the IFFT output corresponding to the second block of data bit stream are pre-pended to the beginning of the second DMT symbol. “Cyclic prefix” carries redundant information that is readily available. However, it provides a “guard interval” that serves as a buffer between two successive DMT symbols. Without the guard interval, the information carried by the 1st DMT symbol will leak to the 2nd DMT symbol and causes interference when the two successive DMT symbols are transmitted over the communication channel 2020, since every practical communication channel has a non-zero length in its impulse response. When we employ a CP whose length is longer than the impulse response of the communication channel, the leakage of the information carried by the first DMT symbol will be contained within the guard interval between the two DMT symbols. The CP, however, is an overhead to the system. A longer CP allows the system to tolerate more dispersion from the communication channel at the cost of a lower data throughput.
FIG. 3 depicts a typical block diagram of a DMT receiver. The received signal from the communication medium 2020 is amplified by an amplifier 2050, then filtered by a filter 2100, and then converted into digital samples by a ADC (analog-digital converter) 2200. As mentioned above, the cyclic prefix in the transmitter needs to be longer then the length of the impulse response of the communication channel 2020 for the leakage of information from one DMT symbol to be contained within the guard interval. In practice, however, the impulse response of the communication channel may be longer than the CP because the CP is usually not long enough (otherwise the sacrifice in data throughput will be prohibitively high). Therefore, a TEQ 2300 (time-domain equalizer) is often used. TEQ 2300 is essentially a FIR (finite impulse response) filter whose purpose is to effectively shorten the impulse response of the communication channel, so that the leakage of the information from one DMT symbol can be contained within the guard interval.
From the output of the TEQ 2300, the frame boundary between two successive DMT symbols is detected. The output of TEQ 2300 is then converted by a SIPO (serial-in parallel-out) 2400 into successive blocks of time domain samples; each block consists of N+N_CP time-domain samples, labeled 0, 1, 2, . . . , N+N_CP−1. The 1st N_CP samples of each frame, which corresponds to the samples within the guard interval, are discarded in the subsequent CP removal 2500, resulting in N samples, labeled 0, 1, 2, to N−1. The N time-domain samples are transformed into N frequency-domain samples by the N-point FFT (fast Fourier transform) 2600.
Ideally, we would like the N frequency-domain samples at the output of FFT 2600 to exactly match the N frequency-domain data at the input of IFFT 1300 in the transmitter depicted by FIG. 1. Unfortunately, due to the communication channel, the amplitude and phase of each sub-carrier is altered and therefore the frequency domain sample at the receiver will not exactly match that at the transmitter. A FEQ (frequency domain equalizer) 2700 is used to equalize the frequency-domain samples. The amplitude and phase change experienced by each tone is thus corrected independently on a per-tone basis. After frequency domain equalization, a slicer 2750 is used to decide the most likely constellation point that the transmitter originally uses for each tone. For example, if the output of FEQ 2700 is 0.9+1.1j for tone number 5 which uses 4-QAM to carry two-bit information, then slicer 2750 would decide that the most likely constellation point that the transmitter originally uses for tone number 5 is 1+1j. The slicer is an example embodiment of a “decision device,” as it makes a most likely decision for each tone. The output from slicer 2750 is mapped to N data banks by the subsequent “QAM demapper” 2800. The N data banks from “QAM demapper” 2800 are converted back into a block of data bit stream by PISO 2900, which ideally will match that of the input of SIPO 1100 in the transmitter.
To effectively shorten the length of the effective impulse response of the communication channel, many algorithms for calculating the coefficients for TEQ are proposed, including minimum mean square error (MMSE), maximum shortening SNR (MSSNR), minimum ISI (mini-ISI), and maximum bit rate (MBR). Among these, MBR offers the best performance, but the computation complexity is too high to be implemented in a commercial MCM receiver. In many cases, unfortunately, none of the algorithms are able to lead to a practical solution that completely contains the leakage of the information from a DMT symbol within the guard interval (i.e. CP of the next DMT symbol). Under these circumstances, a DMT symbol causes interference to the next DMT symbol. This phenomenon is known as “inter-symbol interference” (ISI).
When the length of impulse response of the communication channel exceeds the CP length, a DMT symbol will fail to settle into steady state within its guard interval (i.e. the CP portion of this DMT symbol). In other words, there is still some transient behavior within the “useful” part, i.e. the last N samples, of the DMT symbol. DMT modulation, as a special of MCM, relies on the orthogonality of carriers to faithfully deliver the information. The orthogonality between two tones holds only when they are both in steady state, where both become purely sinusoidal. Whenever the DMT symbol fails to settle into steady state within its guard interval, the orthogonality among tones used by this DMT symbol fails. There is then coupling between the information carried by any two tones that it uses. This phenomenon is known as “inter-carrier interference” (ICI).
One way to alleviate the ISI/ICI problem is to employ multiple TEQ's (time-domain equalizers). For example, a dual-TEQ architecture is shown in FIG. 4. This receiver divides the tones into two groups and employs TEQ1 and TEQ2. Each TEQ is optimized to minimize the ISI/ICI for one group of tones. Each TEQ output is converted into frequency domain samples in a respective FFT. On a per-tone basis, this receiver determines which output of the two paths yields the best SNR (signal-noise ratio). Once the better path for each tone is determined, the output from the path is equalized by the subsequent FEQ. A drawback for this type of receiver is that the hardware cost of this architecture is rather high.
Another way to alleviate the ISI/ICI problem is to employ a so-called “per-tone frequency domain equalizer” (PTFEQ), as shown in FIG. 5. The principle in this design is to eliminate TEQ by replacing it with a plurality of tapped delay lines 5010 in the frequency domain at the FFT output. The hardware cost, however, is prohibitively high for a commercial MCM receiver.
What is needed is a low-cost, robust, and effective scheme for performing ISI/ICI cancellation.